The present invention relates to the field of automatic test equipment for testing electronic signals, and more particularly, to automatic test equipment for analyzing the noise component of an electronic signal.
Automatic test equipment for testing the performance of communications systems, radar systems, and other signal producing devices are known. In this regard, it is often necessary to evaluate the spectral purity of a signal produced by a unit under test (UUT) in order to determine if the UUT is operating within the manufacturer""s or end user""s specifications. Specifically, a manufacturer or end user may specify the maximum phase noise which may be present on a signal produced by the UUT. The phase noise of a signal is a measure of the random phase instability of the signal.
The phase noise of a UUT can be measured in a variety of ways. For example, the output signal of the UUT can be applied directly into the input of a spectrum analyzer which will display the power spectral density of the signal and the phase noise will be visible in the display as random noise power in the spectral plot. Alternatively, the phase noise can be measured using a second signal source as a reference. The second signal source outputs a signal which is identical to or better than the expected UUT signal, but in phase quadrature (if phase modulation noise is being tested) to the UUT signal, i.e., the second signal source is at the same frequency as the UUT signal, but is phase shifted by 90 degrees. The UUT and the second signal source are input into a mixer and, since the two signals have the same carrier frequency, the signals cancel each other out, leaving a signal comprising the combined phase noise of the UUT and the second signal source. In addition, the phase noise may be measured using the Down Converter/Multiple Direct Spectrum Measurement Technique, which is described in U.S. Pat. Nos. 5,337,014 and 5,179,344, the specifications of which are hereby incorporated by reference herein.
Conventionally, when a UUT signal is analyzed, the technician uses a variety of discrete components including a programmable down converter for translating the input signal into a lower, and more easily analyzed, frequency; a narrow FM tunable synthesizer for generating a reference signal, and a separate spectrum analyzer. Since each of these components has its own unique programming requirements, significant time and effort is often spent programming and integrating these discrete components into an effective phase noise measurement system.
With respect to prior art, U.S. Pat. No. 5,508,661 (Keane et al.) describes a fast tuning YIG frequency synthesizer using a fixed oscillator driving a comb line generator to generate a spectrum of comb lines, one of which is selected by a switched array of fixed-tuned, YIG passband filters. The selected comb line is combined in a mixer with a signal from a local oscillator, which, preferably, is a direct digital synthesizer. The output of the mixer is fed to another switched array of fixed-tuned YIG passband filters where only the desired sideband is selected and the comb line and the other sideband is filtered out. In various alternative embodiments, tunable YIG filters are substituted for each array. Some embodiments also use a reverse slope equalizer to break up the coherent energy in the comb line spectrum at the output of the comb line generator to allow RF amplification to be applied without saturating the amplifier.
U.S. Pat. No. 5,770,977 (Uurtamo) describes a microwave frequency synthesizer with ultra-fast frequency settling and very high frequency resolution. The synthesizer purportedly provides the ability to achieve ultra-fast frequency settling times with good frequency resolution and high absolute accuracy over significant bandwidth at microwave frequencies ranging over three octaves. The implementation is an open-loop system requiring little or no compensation of temperature. This is accomplished by providing a frequency doubled direct digital synthesizer output to up/down convert a microwave frequency source. A special tracking filter architecture coupled to the microwave source provides the suppression of unwanted products. Fixed frequency set-on and swept bandwidths in excess of 300 MHz have been demonstrated. This is accomplished by using a direct digitally synthesized quadrature phased carrier which can be set to any frequency within a 350 MHz bandwidth to coherently up/down convert a low phase-noise microwave frequency to the sum or the difference frequency product. Individual control of differential phase and amplitude over frequency assures very high suppression of unwanted products without the use of additional filtering.
U.S. Pat. No. 5,053,714 (Durand) describes a measuring circuit for the additive phase noise characteristic of a component in the vicinity of a carrier frequency. The measuring circuit is constructed of a central channel and two side channels. Each of these channels contains a model of the component to be characterized. Two phase detecting circuits are employed in which each processes an input signal from one of the side channels with an input signal from the central channel to generate signals which represent phase deviations between the two input signals. An intercorrelation circuit then utilizes the outputs from these phase detecting circuits to determine the characteristic additive phase noise of the component to be characterized by eliminating any additive phase noise superadded by other measuring circuit elements or induced by outside disturbances.
U.S. Pat. No. 5,412,325 (Meyers) describes a phase noise measurement system and method in which three independent signal sources are used to statistically derive the power spectral density of the phase noise content of signals from each of them. This is accomplished by mixing each of the signals two at a time (i.e., signal one with signal two, signal one with signal three, and signal two with signal three) and capturing the resultant difference signals, such as with a waveform recorder, for example. A servo electronics loop is used to remove the carrier and any long term signal drift from the resultant difference signals. Statistical analysis is then used to compute the composite power spectral densities of the resultant difference signals, and to solve for the individual power spectral densities of the original signals. The system and method uses the mathematical relationships between the three sources that have similar magnitudes of phase noise, to compute the power spectral density of the noise content of signals from each source. The system and method purportedly requires a minimum of interconnect hardware and only three inexpensive waveform recorders. Furthermore, the size, weight, and cost of producing the present phase noise test system is said to be relatively low.
U.S. Pat. No. 5,608,331 (Newberg et al.) describes a noise measurement test system for making phase noise and amplitude noise measurements of microwave signals derived from a continuous wave RF source. The system comprises an RF input for receiving an applied RF noise signal and an RF coupler coupled to the RF input for splitting the applied RF noise signal into first and second paths. A mixer that comprises a synchronous phase detector is coupled to receive signals from the first and second paths and which outputs demodulated phase noise. The first path comprises a variable attenuator and a variable phase shifter coupled between the coupler and the first input of the mixer for providing a reference signal input to the synchronous detector. The second path comprises a delay line and an adjustable RF carrier nulling circuit coupled between the coupler and the mixer. A video amplifier is coupled to an output of the mixer for providing a baseband video output signal from the system that is processed to produce noise data.
U.S. Pat. No. 4,748,399 (Caldwell et al.) describes a multichannel phase noise measurement system which is purportedly capable of measuring the phase noise of two signal source operating between 100 MHz and 24 GHz. The system has two input channels and a mixer which produces a difference frequency between the signals received in the input channels. The difference frequency containing the phase noise is frequency multiplied and amplified to raise the phase noise level 12 dB. The multiplied and amplified difference signal is phase locked to a commercially available frequency synthesizer with inferior noise characteristics to the units being tested. By amplifying the phase noise of the original signals above the phase noise levels of the commercially available frequency synthesizer, they become measurable.
In accordance with the present invention, an integrated phase noise measurement system is provided which includes a low noise synthesizer module, a receiver/downconverter module, a controller, a digitizer, and a spectrum analyzer. The low noise synthesizer and receiver/downconverter may be used separately, or in combination with one another.
The low noise synthesizer module may be used in a phase noise measurement system to produce, for example, spectrally pure L-Band signals. In phase noise measurement systems, it is important that the synthesizer module, which provides the reference signal to which the UUT is compared, produces an extremely low noise signal. This is important because the phase noise measurement system will be unable to accurately measure noise in the signal produced by the UUT which is below the noise level of the synthesizer. As a result, the noise level of the synthesizer sets a xe2x80x9cnoise floorxe2x80x9d below which noise measurements cannot be made. In addition, it is sometimes desirable to measure the noise of a signal xe2x80x9cfar outxe2x80x9d as well as xe2x80x9cclose-inxe2x80x9d from the carrier frequency of the UUT. Conventional low noise synthesizers, however, are normally unable to provide a reference signal which exhibits low noise both xe2x80x9cfar outxe2x80x9d as well as xe2x80x9cclose-inxe2x80x9d from the carrier frequency.
In accordance with the present invention, a low noise synthesizer is provided which produces an output signal with a low (phase and amplitude) noise characteristic close in to the carrier frequency, as well as low noise far out from the carrier frequency, by utilizing a low noise oscillator coupled to a comb generator to provide a signal with low noise close in, and a signal acoustic wave oscillator to provide a signal with low noise far out. The low noise synthesizer includes a low noise crystal oscillator for producing a signal having a frequency (preferably 120 MHz), and a surface acoustic wave oscillator for producing a signal having a second frequency (preferably 960 MHz). A comb generator is coupled to an output of the crystal oscillator, and a bandpass filter is coupled to the output of the comb generator. The bandpass filter has a passband which includes the second frequency (and preferably has a passband centered at 960 MHz). A frequency dividing component is coupled to the low noise crystal oscillator to selectively produce one of a plurality of offset frequencies, each of the plurality of offset frequencies being in a first frequency range (which is preferably from 10-40 MHz). A mixer has a first input coupled to an output of the frequency dividing component, and has a second input which is selectively coupled to either the output of the surface acoustic wave oscillator or the output of the bandpass filter. A tunable bandpass filter is coupled to an output of the mixer, and is selectively tuned to a passband which includes a sum (or difference) of the selected offset frequency and the second frequency. The output of the tunable bandpass filter provides an output signal with low noise close in when the output of the bandpass filter is coupled to the mixer, and an output signal with low noise far out when the surface acoustic wave oscillator is coupled to the mixer.
In accordance with the preferred embodiment of the low noise synthesizer, an low noise oscillator produces a 120 MHz reference signal which is multiplied to 960 MHz by a comb generator to provide a signal with low noise close in, (e.g., within 400 kHz of the carrier frequency) while a surface acoustic wave oscillator is utilized to produce a 960 MHz signal to provide a signal with low noise far out (e.g.,  greater than carrier frequency+400 kHz,  less than carrier frequencyxe2x88x92400 kHz) but relatively high noise within 400 kHz of the carrier frequency. In accordance with this preferred construction, a noise component of less than 100 dBc is achieved at 100 Hz,  less than 120 dBc is achieved at 1 kHz,  less than 130 dBc is achieved at 10 kHz,  less than 140 dBc is achieved at 100 kHz and  less than 160 dBc is achieved at  greater than =400 kHz.
The receiver/downconverter in accordance with the present invention is preferably used in conjunction with the low noise synthesizer described above, but may also be used with conventional synthesizers. In accordance with the present invention, the receiver/downconverter can be used to perform absolute phase noise measurement, additive phase noise measurement, and down converter/multiple direct (DND) phase noise measurement.
The receiver/downconverter includes a UUT input, a synthesizer input, an output, a first mixer, a second mixer, a delay element coupled to a first phase shifter, a phase locked loop (PLL) coupled to a second phase shifter, a first bandpass filter, a second band pass filter, a first low pass filter, a second low pass filter, a comb generator, and an amplifier. The synthesizer input is coupled to one input of the first mixer. The UUT input is coupled directly through to the other input of the first mixer for absolute phase noise measurement, and for DMD phase noise measurement. For an additive phase noise measurement, the UUT input is coupled through the delay element and first phase shifter before being applied to the other input of the first mixer. The output of the first mixer is selectively coupled to one of the first bandpass filter, the second bandpass filter, and the first low pass filter. The output of the second bandpass filter is coupled to the comb generator, and the output of the first bandpass filter is coupled to the PLL and to one input of the second mixer. The PLL is coupled to the second phase shifter, which, in turn, is coupled to the second input of the second mixer. The output of the second mixer is coupled to the second low pass filter. Finally, the outputs of the first low pass filter, the second lowpass filter, and the comb filter, are selectively coupled to the amplifier. The amplifier is coupled to the output of the receiver/downconverter.
In order to perform an absolute phase noise measurement with the receiver/downconverter, a synthesizer signal is generated by the synthesizer which is offset by a first offset frequency (preferably 10 MHz) from the expected UUT signal frequency. The synthesizer signal and the UUT signal are applied to the first mixer to generate an IF signal (preferably 10 MHz IF), which is then coupled through to the first bandpass filter (10 MHz BP for 10 MHz IF signal) to isolate the IF signal. The IF signal is then coupled to both i) the input of the PLL, and ii) one input of the second mixer. The output of the PLL is then passed through the second phase shifter prior to being applied to the other input of the second mixer. The PLL-Phase shifter circuit maintains phase quadrature between the two inputs of the second mixer. Since the two inputs to the second mixer have the same frequency, the mixer will output a 0 MHz difference signal to the second low pass filter (preferably 1.9 MHz low pass). The signal output from the second low pass filter, which comprises the absolute noise signal, is then passed through to the amplifier and then output to a spectrum analyzer via the output of the receiver/downconverter.
In order to perform an additive phase noise measurement with the receiver/downconverter, a synthesizer signal is generated by the synthesizer which is equal in frequency to the expected UUT signal. The UUT signal is passed through the delay line and the first phase shifter before being applied to one input of the first mixer. The first phase shifter maintains phase quadrature between the two inputs to the first mixer. Since the signals input to the first mixer are of equal frequency, the first mixer will output a 0 MHz difference signal to the first low pass filter (preferably 1.9 MHz low pass). The signal output from the second low pass filter, which comprises the additive phase noise signal, is then passed through to the amplifier, and output to the spectrum analyzer via the output of the receiver/downconverter.
In order to use the receiver/downconverter to down convert and multiply the UUT signal to microwave frequencies (DMD phase noise measurement) which can be directly measured on a spectrum analyzer as a double side band signal, a synthesizer signal is generated by the synthesizer which is offset by a second offset frequency (preferably 100 MHz) from the expected UUT signal frequency. The synthesizer signal and the UUT signal are applied to the first mixer to generate a second IF signal (preferably 100 MHz IF), which is then coupled through the third bandpass filter (preferably 100 MHz), and into the comb generator. As one of ordinary skill in the art will appreciate, the comb generator produces a spectrum of signals which are multiples of the input frequency signal. The output of the comb generator is then passed through the amplifier and on to the spectrum analyzer. The user can then choose which multiple of the second IF frequency he or she wishes to view. Preferably, a signal in the 4-6 GHz range is chosen because that is the most effective range of most spectrum analyzers.
In accordance with another embodiment of the present invention, the receiver/downconverter and the low noise synthesizer are incorporated into an integrated phase noise measurement system including a spectrum analyzer, a digitizer, and a controller. The controller may include, for example, a computer, a display screen, and a keyboard.
In order to measure absolute phase noise, an operator specifies the carrier frequency of the UUT, and indicates whether the signal to be measured is AM (amplitude modulated) or PM (phase modulated). This information is input to the controller via the keyboard, input program, or other input device. The controller automatically configures the receiver/downconverter to receive a 10 MHz IF. This 10 MHz signal is mixed with the PLL 10 MHz signal and a residual error frequency is measured on the digitizer. After correcting this residual error by automatically reprogramming the frequency difference to account for this error, AM or PM noise is measured. At this point, when the PLL and 10 MHz IF signals are identical in frequency, the phase locked PLL and 10 MHz IF signals are phase detected and the second phase shifter is rotated through zero crossings and peak amplitudes to establish the beat note amplitude and to establish a quadrature setting for PM or peak setting for AM. This determinant also provides the DC beat note level from the phase detector. If the measurement is being made on an AM (amplitude modulated) signal, then the peak signal value is stored as the beat note (PM signals will have a constant amplitude). The beat note represents the total RF power after phase detector conversion loss, which represents the signal level that the noise spectrum is referenced to. The controller then switches the output of the amplifier of the receiver/downconverter through to the spectrum analyzer. The power spectral density and spurious is then measured by the spectrum analyzer. Then, in order to establish true power spectral density, the controller applies 55 dB, 3 dB tangential, 3 dB single sideband, and 2.5 dB Filter/log Fidelity corrections to the measured noise power. Finally, the controller rechecks the quadrature (for PM measurement) or peak (for AM measurement) setting to ensure that quadrature or peak was not lost during the foregoing measurement.
In order to measure additive phase noise, an operator specifies the carrier frequency of the UUT, and indicates whether the signal to be measured is AM (amplitude modulated) or PM (phase modulated). This information is input to the controller via the keyboard, input program, or other input device. The controller automatically configures the receiver/downconverter to provide an additive phase noise measurement, and couples the output of the first phase shifter of the receiver downconverter to the digitizer. The controller then rotates the first phase shifter until quadrature is established and stores the phase snifter value. The output from the first phase shifter is then digitized by the digitizer and transmitted to the controller for processing. The controller processes the digitized data to determine the beat note of the UUT signal. If the measurement is being made on an AM (amplitude modulated) signal, then first phase shifter is rotated until peak is established, and the peak signal value is stored as the beat note (PM signals will have a constant amplitude). The controller then switches the output of the amplifier of the receiver/downconverter through to the spectrum analyzer. The power spectral density and spurious is then measured by the spectrum analyzer. Then, in order to establish true power spectral density, the controller applies 55 dB, 3 dB tangential, 3 dB single sideband, and 2.5 dB Filter/log Fidelity corrections to the measured phase power. Finally, the controller rechecks the quadrature or peak setting to insure that quadrature or peak was not lost during the foregoing measurement.